Impedance of a finite insulated antenna in a cold plasma with a perpendicular magnetic field

A variational formulation is developed for the impedance of a finite strip antenna embedded in a planar dielectric slab which is surrounded by a magnetoionic medium (cold electron plasma) with a static magnetic field impressed in a direction perpendicular to the antenna surface. Closed form expressions are obtained in the limit of low frequencies, and for a short antenna in a uniaxial medium. The impedance becomes large at the plasma frequency, near the upper hybrid resonance frequency, and further resonances are observed near the gyro-frequency if the gyro-frequency exceeds the plasma frequency. The reactance of a short antenna is inductive at low frequencies, but becomes capacitive as the thickness of the insulation around the antenna is increased. For very thin insulating layers the wave number of the variationally approximated current distribution exceedssqrt{sqrt{epsilon_{1}} sqrt{epsilon_{3}}} k_{0}(epsilon_{1}andepsilon_{3}are the two diagonal elements of the permittivity matrix), whereepsilon_{1}andepsilon_{3}may have positive or negative real parts. However, this approximation does not apply to current distributions along an insulated antenna. The present calculations are also compared with earlier work on antenna impedances.     

On the relationship between the transmitting and receiving properties of an antenna

The response of a linear, passive antenna in free space to an incident plane wave is related to its far-field radiation in the reverse direction bybar{h}^{r}=bar{h}^{t}. The vector effective heightbar{h}^{r}characterizes the voltage across the open-circuit antenna terminals induced by an incident plane-wave field, and the correspondingbar{h}^{t}characterizes the far-field radiation of the antenna when driven by a current injected into the same terminals. This relationship, which is shown to follow from the well-known reciprocity principle, both includes phase and makes mathematically explicit the polarization relation involved in the identity of the transmit and receive patterns of an antenna.     

Synthesis of linear antenna arrays

Synthesis of antenna arrays employing theL_{2}-norm as well as theL_{infty}-norm is discussed. The approximation in theL_{infty}-norm is obtained making use of Lawson's algorithm. A general iterative perturbation technique has been evolved for pattern synthesis for the case when the antenna currents alone are varied as web as for the case when both the antenna currents and the element positions are simultaneously varied. A few illustrative examples are given. The convergence of the iteration and the uniqueness of the solution are discussed.     

The performance of an iris-loaded planar phased-array antenna of rectangular waveguides with an external dielectric sheet

A new method for calculating the performance of an iris-loaded planar phased-array antenna of rectangular waveguides is presented. The method is based upon the expansion of the tangential electric aperture field in terms of judiciously chosen functions. Also, the influence of a dielectric sheet in front of an antenna the apertures of which are loaded with inductive irises is investigated. In the experiments, a sheet with a low relative permittivity (epsilon_{r} = 2.3orepsilon_{r} = 3.5) is spaced in front of a space-fed planar antenna of some 850 radiating elements, and the total transmitted power of the antenna is measured as a function of frequency and scan angle. The results indicate, that both in theory and in practice, an excellent match can be achieved with a polythene sheet (epsilon_{r} = 2.3) or a Perspex sheet (epsilon_{r} = 3.5).     

Loaded horizontal antenna over an imperfect ground

The Fresnel reflection coefficient technique is employed to establish anE-field integral equation for the antenna current. A resistive loading of the formLambda(x) = Lambda_{0}/(1 - |x|/L)is used to load the antenna. An optimization technique is discussed for determining the value of critical loadingLambda_{0}^{c}, which enforces a traveling wave current on the antenna. Results are given for the critical loading parameters, antenna currents, input impedances and radiation patterns versus different antenna dimensions and ground permittivities and conductivities. Some representative time-domain results for such loaded antennas are also included.     

Fundamental superstrate (cover) effects on printed circuit antennas

The fundamental effects of superstrate (cover) materials on printed circuits antennas are investigated. Substrate-superstrate resonance conditions are established which maximize antenna gain, radiation resistance, and radiation efficiency. Criteria are determined for material properties and dimensions for which surface waves are eliminated and a radiation efficiency due to substrate-superstrate effects ofe_{s} = 100percent is obtained. Criteria for nearly omnidirectionalbar{H}-plane patterns and nearly omnidirctionalbar{E}-plane patterns are presented. Finally, a general criterion is given for choosing a superstrate to optimize efficiency for the important case of nonmagnetic layers with the antenna at the interface.     

Antenna radiation characteristics with partially coherent illumination

The radiation characteristics of a partially coherent illuminated antenna are found for both a circular aperture and a linear antenna. The antenna illumination function is assumed to be Gaussian, and the phase structure function in the plane of the antenna is assumed to be of the formcr^{nu}, withnu = 1, 5/3, and 2. The beam broadening and reduction in gain in decibels are found to be of the formKb^{alpha}, whereKandalphaare functions of the antenna illumination and the "law" of the phase structure function, andbis a function of the "level" of the phase structure function.     

Microstrip disk antennas, Part I: Efficiency of space wave launching

A theoretical study on the problem of surface wave excitation of microstrip disk antennas is presented. The cavity model with magnetic sidewalls and dyadic Green's functions in stratified media is used to obtain the radiated fields in an integral form. Lossless media are assumed such that there is no cross coupling between the powers in the space (P_{SP}) and surface (P_{SU}) waves. The separate contributions of these two powers is examined. With the assumption thatP_{SU}does not contribute to the main radiation patterns of the antenna an efficiency of space wave launching and a corresponding antenna directivity are defined. Values of efficiency and directivity as functions of the dimensions of the antenna and for two values of dielectric constants of the substrate are shown. Agreement of these results with some of the available data is observed. It is worth noting thatP_{SU}may correspond to an appreciable portion of the total radiated power.     

Unbalanced-Mode Spiral Antenna Backed by an Extremely Shallow Cavity

This paper describes a two-arm Archimedean spiral antenna backed by a conducting cavity, where only one arm is directly excited, with the other arm being parasitically excited; in other words, the spiral arms are excited in an unbalanced mode. A balun circuit required for a conventional two-arm spiral is not used for this unbalanced-mode spiral. The design of the unbalanced-mode spiral is performed over a frequency range of $f _{rm Ld} = 3~{rm GHz}$ to $f _{rm Hd} = 9~{rm GHz}$ (1:3 bandwidth), where the antenna height is selected to be extremely small ($7~ {rm mm}= 0.07$ wavelength at $f _{rm Ld}$) to realize a low-profile antenna. For reference, a corresponding spiral antenna excited in balanced mode is also analyzed. It is found that the unbalanced-mode spiral shows an acceptably small VSWR over the design frequency range of $f _{rm Ld}$ to $f _{rm Hd}$. The radiation is circularly polarized around the antenna axis normal to the spiral plane. The gain shows behavior similar to that of the balanced-mode spiral. Results for other antenna heights (5 mm, 10.5 mm, and 14 mm) are also presented and briefly discussed. It can be said that the unbalanced-mode spiral is a circularly polarized wideband antenna with a simple feed system.      

Radiation from a gyro-plasma sheathed aperature

It is shown that the presence of a gyrotropic plasma sheath covering a rectangular slot radiator may not alter significantly the radiation pattern when the static magnetic field is perpendicular to the slot. The plasma covered antenna remains omnidirectional and the radiated power is not significantly reduced when the cyclotron frequencyomega_{b}, the collision frequencynu, and the operating frequencyomega, fulfill the inequalityomega_{b}^{2}ggomega^{2}+nu^{2}. It is also found that the matching between the plasma covered antenna and its surrounding is sharply improved for certain combinations of the plasma sheath parameters and the operating frequency. This finding may have an application in plasma diagnostics, the slab widthLand the plasma frequencyomega_{p}can be determined by varyingomega_{i}andomegaaccording to an expression derived in this paper.     

Current distribution and input admittance of an infinite cylindrical antenna in anisotropic plasma

The current distribution and input admittance of an infinitely long cylindrical antenna driven by a slice generator and immersed in an anisotropic plasma are investigated. The applied dc magnetic field is along the axis of the antenna. By superimposing the characteristic waves guided along the antenna, the current solution is obtained in the form of a one-dimensional integral, which is examined both analytically and numerically. WhenK_{perp} > 0, the magnitude of the current decays slowly with the distance from the source, and its phase is nearly linear with a "propagation constant" equal tosqrt{K_{perp}}k_{0}for an antenna with very small radius. WhenK_{perp} < 0, the magnitude of the current decreases rapidly away from the source, and its phase is no longer linear. The input admittance decreases with the radius of the antenna.     

Comments on the design of log-periodic dipole antennas

The design procedure of the log-periodic dipole (LPD) antennas is discussed. The results of the calculations carried out by the use of the more recent and accurate methods of the analysis for the arrays of unequal dipoles, are compared with those already available in the literature. The calculations refer to the gain of LPD antenna as a function ofZ_{0}, the characteristic impedance of the exciting transmission line, ofh/a, the ratio between the half-length and the radius of dipoles, oftauandsigma, two parameters which describe the geometry of the LPD antenna. The values of gain are lower than those given by earlier calculations. The degradation of the gain is more consistent, when the values either ofZ_{0}orh/aare increased with respect toZ_{0} = 100 Omegaandh/a = 125, which are parameters often used in the literature. The many diagrams included in this paper for different values oftau, sigma, Z_{0}, andh/acan be useful in a more accurate design of an LPD antenna. Confirmation of the computer data has been obtained by the tests performed on various scale models and on a full scale antenna.     

Criteria for nearly omnidirectional radiation patterns for printed antennas

Radiation from printed antennas is investigated with emphasis placed on producingbar{E}- andbar{H}-plane radiation patterns that are as nearly omnidirectional as possible. This is achieved using criteria which are derived for a nonzero radiation field extending down to the layer surface (radiation into the horizon). It is determined that this phenomenon arises when a surface wave pole coincides with a branch point in the complex plane. A simple ray optics interpretation is given for the phenomenon, and graphs are presented to easily enable design of printed antenna geometry to achieve nearly omnidirectionalbar{E}- orbar{H}-plane patterns.     

A technique to estimate the relative gains, tuning phases, and RE phase shifts of the elements of a phased-array radar

An in-situ technique to estimate the following parameters of a phased-array antenna is described: 1) the relative array-element excitation voltages, 2) the array-element tuning phases, and 3) the RF phase shifts at the array elements. This technique has several significant features. First, it involves the use of two auxiliary antennas. One is a remote continuous wave (CW) source directed at the phased-array antenna. The other is a passive antenna mounted close to the phased-array antenna. Its output is used to produce a reference phase for phase measurements. Second, it contains a technique to reduce the errors in phase estimates. Third, it takes note that beam steering uses phase sums of the form (phi_{k} + beta_{kl}), wherephi_{k}is the tuning phase for thekth array elemenet andbeta_{kl}, is an RF phase shift of the array element, and pays special attention to reduce the errors associated with the estimates of these sums. Fourth, it assumes the use of a reasonably stable and strong CW source of commerically available quality. No other assumptions are made. Experimental results obtained with a 295-elementS-band space-fed phased-array antenna are given.     

Interferometric measurement of tropospheric phase fluctuations at 22 GHz on antenna spacings of 27 to 540 m

Phase fluctuations at 22 GHz introduced by tropospheric events were measured by observing cosmic compact radio sources with the Nobeyama millimeter-wave array on antenna spacings of 27 to 540 m. The observed interferometer phases were used to derive Allan standard deviationssigma_{A}of phase fluctuations. The standard deviationssigma_{A}had good relations to antenna spacingLas a function,sigma_{A} =A times L^{N} + B. The exponentNderived from the observations in February distributed around 5/6 as expected from the Kolmogorov turbulence model, but was smaller than 5/6 from the observations in May. Time domain spectra ofsigma_{A}also imply that atmospheric structures (such as moist air, clouds, precipitation, etc.) have seasonal variations. Meteorological parameters monitored at the height of 20 m did not show good correlations tosigma_{A}.     

Theoretical and experimental study of a finite cylindrical antenna in a plasma column

A numerical technique has been used to solve the integral equation determining the current distribution on a finite cylindrical antenna in a long plasma column; a "cold" isotropic plasma model with a relative dielectric constantepsilon_{r} = 1 - omega_{p}^{2}/omega^{2}is used in the analysis. Collision losses and inhomogeneities in the plasma have been ignored for the sake of mathematical simplicity. It is found that when0 < epsilon_{r} < 1, the plasma column tends to behave like a dielectric and reduces the effective electrical length of the antenna. The current distribution is still somewhat sinusoidal but with a longer wavelength. In the range-1 < epsilon_{r} < 0, the antenna characteristics undergo a drastic change; the current decays very rapidly along the antenna and, for antennas longer than a quarter wavelength, the input admittance remains almost constant and independent of the length. Extensive experimental measurements have been made to verify the theory. The plasma sheath surrounding the antenna was produced by an active, hot.cathode, helium discharge; the plasma diagnostic measurements were made by using Langmuir probes and cavity perturbation techniques. The theoretical and experimental results show a significant degree of qualitative agreement.     

The small bare loop antenna immersed in a dissipative medium

The input admittance of a small thin-wire circular loop antenna, driven by a slice generator, immersed in a dissipative medium, is considered. It is found that the solution given by Storer for the loop antenna in a lossless medium can be carded over readily by replacingzeta_{0}byzeta, andk_{0}byk. The numerical values of the normalized input conductance and input susceptance of a small loop antenna, namelybeta b leq 0.3,Omega = 10, are calculated. It is to be noted that the input susceptance is practically independent ofkwhile the input conductance changes as much as seventeen times in this special case.     

An experimental study of the insulated dipole antenna immersed in a conducting medium

The driving-point admittance and the amplitude and phase distributions of the current referred to the driving-point were measured for an insulated cylindrical antenna immersed in a conducting medium. The ratiosigma/omegaepsilon_{r}epsilon_{0}of the conducting medium was varied fromsigma/omegaepsilon_{r}epsilon_{0}= 0.036to 8.8, a range which includes a variety of media such as poor insulators, the ionosphere, plasmas, dry earth, wet earth, lake water and sea water. The antenna heightbeta hin radians was varied frombeta h=0.1throughbeta h=2piat intervals of 0.1. The thickness of the insulator was varied fromb/a = 1.25tob/a = 12.0whereais the radius of the antenna andbthe radius of the insulator. Measurements have also been made of the admittance, current and phase distributions along an insulated antenna with a conductive top load, that is, one whose tip is in direct contact with the conducting medium. It is found that when the tip of the antenna is in direct contact with the conducting medium, the current increases almost linearly as the end of the antenna is approached. This is quite unlike the decaying sinusoidal distribution on the completely insulated antenna. In a general sense, the experimental results are in fair agreement with an approximate theoretical expression for the admittance of an insulated antenna immersed in a highly conducting solution. The approximate theory is based on the driving-point admittance of a coaxial line whose outer conductor is imperfectly conducting and infinite in extent.     

Numerical solution of nonuniform surface wave antennas

An integral equation is used to numerically study microwave surface wave antennas. The source is chosen as a horizontal magnetic line current which is placed parallel to a planar surface having a one-dimensional nonuniform surface reactance. The integral equation is numerically solved exactly taking full account of radiation from both the feed and the termination, and their mutual interaction. By varying the profile of the surface reactance and the antenna length, radiation characteristics such as directivity, beam width and sidelobe level are discussed. The optimum excess phase delay is found to be aboutpi/3for an antenna length of3lambda_{0}, gradually increasing topi/2for a length of8lambda_{0}. When a feed transition in the reactance profile is used to achieve a good surface wave launching efficiency near the source, the first sidelobe can be reduced to less than-10 dB below the main beam level with little effect upon the directivity. Optimum reactance profiles are suggested for antennas from3lambda_{0}to8lambda_{0}long.     

The wide-band matching area for a small antenna

A special frequency scale is proposed for use in describing the fundamental limitation on wide-band matching of a small antenna by means of a fixed reactive network. This scale isu = - 1/omega^{3}, normalized to unit frequency (omega_{1} = 1)-It has unit width for the high-pass band above this frequency. In terms of the familiar matching exponentmu = ln 1/|rho|, the matching area over this scale is limited to3 pi p_{1}, whereP_{1}is the (small) radiation power factor atomega_{1}. The corresponding limit on matching efficiency is6 pi p_{1}. A double-tuned matching network can approach2/pithis area over any bandwidth on this scale.