Image-rejection receiver using on-chip active notch filters for 2.4-GHz band wireless communication applications

This paper presents the design and experimental results of image-rejection (IR) receiver front-end for 2.4-GHz band applications. The proposed IR-receiver front-end integrates a third-order active notch filter into each of conventional cascode low noise amplifier and down-conversion mixer to achieve high image-rejection ratio (IRR). The image signal is suppressed and the wanted signal is maximized due to series and parallel resonator effects of the notch filter, respectively. Consequently, the proposed IR-receiver front-end implemented in a standard 0.18 μm CMOS technology has the power gain of 21.5 dB, the noise figure of 3.5 dB, the input third order intermodulation product of ?15 dBm and the IRR of 56 dB. The IR-receiver front-end dissipates a total of 5.5 mA from supply voltage of 1.8 V.     

Design and analysis of high speed capacitive pipeline DACs

Design of a high speed capacitive digital-to-analog converter (SC DAC) is presented for 65 nm CMOS technology. SC pipeline architecture is used followed by an output driver. For GHz frequency operation with output voltage swing suitable for wireless applications (300 mV_pp) the DAC performance is shown to be limited by the clock feed-through and settling effects in the SC array rather than by the capacitor mismatch or kT/C noise, which appear negligible in this application. While it is possible to design a highly linear output driver with HD3 < ?70 dB and HD2 < ?90 dB over 0.5–5 GHz band as we show, the maximum SFDR of the SC DAC is 45 dB with 8-bit resolution and Nyquist sampling of 3 GHz. The capacitor array is designed based on the DAC design area defined in terms of the switch size and unit capacitance value. A tradeoff between the DAC bandwidth and resolution accompanied by SFDR is demonstrated. High linearity of the output driver is attained by a combination of two techniques, the derivative superposition (DS) and resistive source degeneration. In simulations the complete DAC achieves SFDR of 45 dB with 8-bit resolution for signal bandwidth 1.36 GHz with Nyquist sampling. With 6-bit and 5.5 GHz bandwidth 33 dB SFDR is attained. The total power consumption of the SC DAC is 90 mW with 1.2 V supply and clock frequency of 3 GHz.     

Tunable wideband SAW-less receiver front-end in 65 nm CMOS

A surface acoustic wave-less receiver front-end for GSM, TD-LTE and TD-SCDMA standards featuring a novel low noise amplifier (LNA) architecture and harmonic rejection technique is presented. The two-stage LNA uses capacitive feedback in the first stage for wideband input matching. It can operate from 500 MHz up to 2.5 GHz with an S₁₁ below ?15 dB. The harmonic rejection mixer structure operates using two- and four-phase local oscillator signals and is capable of achieving a high harmonic rejection over a wide channel bandwidth. The average harmonic rejection is above 60 dB measured over a 20 MHz LTE channel and above 70 dB over a GSM channel. The mixer structure contains digitally tunable resistor and capacitor banks for precise tuning, causing the peak harmonic rejection in the channel to exceed 80 dB. The precise tuning capability ensures good harmonic rejection in the presence of process mismatch and duty cycle mismatch. The 1-dB received signal compression point with a blocker present at 20/80 MHz offset for low-/high-band is 0 and +2 dBm, respectively. In-band IIP3, and IIP2 are ?14.8 and >49 dBm, respectively, for low-band. For high-band they are ?18.2 and >44 dBm. Implemented in 65 nm CMOS, the complete front-end consumes 80 mW of power.     

Ka Band Broadband Fourth Harmonic Image Rejection Mixer

This paper presents the design and implementation of Ka band broadband hybrid integrated image rejection mixer with a fourth harmonic mixer as unit mixer. Detailed design and analysis have been carried out. The mixer was fabricated by hybrid microwave integrated circuit (HMIC) process based on the thin film ceramic substrate which can reduce the cost compared to monolithic microwave integrated circuit (MMIC). The measured results showed conversion loss less than 11.2 dB and image rejection ratio (IRR) more than 20 dB in 4 GHz RF bandwidth. It can also play the role of up-converter from the test data.     

High Performance Millimeter‐Wave Image Reject Low‐Noise Amplifier Using Inter‐stage Tunable Resonators

A Q‐band pHEMT image‐rejection low‐noise amplifier (IR‐LNA) is presented using inter‐stage tunable resonators. The inter‐stage L‐C resonators can maximize an image rejection by functioning as inter‐stage matching circuits at an operating frequency (F_OP) and short circuits at an image frequency (F_IM). In addition, it also brings more wideband image rejection than conventional notch filters. Moreover, tunable varactors in L‐C resonators not only compensate for the mismatch of an image frequency induced by the process variation or model error but can also change the image frequency according to a required RF frequency. The implemented pHEMT IR‐LNA shows 54.3 dB maximum image rejection ratio (IRR). By changing the varactor bias, the image frequency shifts from 27 GHz to 37 GHz with over 40 dB IRR, a 19.1 dB to 17.6 dB peak gain, and 3.2 dB to 4.3 dB noise figure. To the best of the authors' knowledge, it shows the highest IRR and F_IM/F_OP of the reported millimeter/quasi‐millimeter wave IR‐LNAs.     

PN-assisted digital background calibration of two-step ADC to over 14-bit accuracy

A new non-binary multiplying digital-to-analog converter (MDAC) structure with signal-dependent dithering scaling technique is proposed in this paper. A full digital background calibration algorithm based on pseudo-random dithers injection is used to calibrate the nonlinear errors of MDAC. By measuring sampling capacitor mismatch and op-amp gain errors of the pipelined analog-to-digital converter (ADC) in background, the errors will be greatly reduced by the proposed calibration algorithm. At the same time, the signal-dependent dithering scaling technique provides a swing margin to the injected pseudo-random signal. By using this technique, the errors caused by the capacitor mismatch and op-amp gain errors can be calibrated at the same time. What’s more, this method greatly accelerates the convergence speed. A two-stage 14-bit pipelined ADC is used to simulate and verify the proposed algorithm. The simulation results indicate the effectiveness of the technique, in which the signal-to-noise plus distortion (SNDR) and the spurious-free dynamic range (SFDR) performance of a 14-bit two-step ADC are improved from 49.12 and 56.25 to 85.68 and 102.23 dB with the input frequency being 0.06 * f _s , respectively. The SFDR is more than 98 dB. The SNDR reaches 84 dB in the whole Nyquist bandwidth after calibration. Integral nonlinearity is improved from 80 to 1.5 least significant bits after calibration.     

Design of Weaver topology

A novel design based on Weaver topology is proposed. Studies showed that the image-rejection ratio (IRR) performance of the design is insensitive to the phase errors of the LO₁ and the IRR against gain mismatch is inherently about 6 dB superior to the conventional Weaver topology. The design greatly increases the practicality of the Weaver receiver     

The self-calibration method of IR mixer with low IF

This paper describes a new self-calibration method of phase shifters providing better suppression of image component in the signal path of a receiver for ISM 2.4 GHz band. The method has been applied to Hartley architecture of image-reject mixers that are used in low IF RF receivers. The method relies on measuring the amplitude differences and phase errors between the I/Q mixer signals and iterative tuning of digitally controlled capacitance arrays, which are located inside the phase shifters. Circuit implementation of the proposed method in 0.35 μm BiCMOS provided IRR (Image Rejection Ratio) = 45 dB what is about 20 dB better than IRR obtainable without self-calibration. The range of amplitude compensation is 20% and the phase correction operates correctly in the range of initial phase error ±4°.     

A Method to Reduce Sampling Rate of the ADC in Feedback Channel for Wideband Digital Predistortion

This paper presents a new digital predistortion (DPD) solution for wideband signals with low feedback sampling rate. To reduce the minimum sampling rate of the analog-to-digital converter (ADC) for wideband digital predistortion, the proposed method uses a bandpass filter to form a narrowband signal before the ADC. Then, a deconvolution operation is performed to recover the original wideband signal from the ADC samples. The proposed method is evaluated with an international mobile telecommunication-advanced signal with 100 MHz bandwidth. The simulation results show that the recovered signal of the proposed method closely approximates to the original signal in the passband of the filter, and the mean square error of the deconvolution decreases as the signal-to-noise ratio increases. The proposed algorithm can reduce the sampling rate of the ADC from 1105.92 million samples per second (MSPS) to 368.64 MSPS, and improve the adjacent channel power ratio more than 20 dB, which is merely 5.6 dB less than the conventional DPD with 1105.92 MSPS sampling rate.     

一款应用于GPS接收机的抗失配零中频混频器

为了解决单正交零中频混频器I/Q失配造成的影响,提高自身镜像抑制能力。基于电位混频原理采用CMOS 0.18μm工艺设计出一款工作在1.575 42GHz双正交结构的抗失调零中频混频器,通过添加四个电容构成MOSFET-C低通滤波器以及两个在低频段工作的运放构成的输出放大级,射频输入信号能够得到有效处理。测试结果表明该结构在镜像抑制能力上比传统结构改善了6dB左右,电路采用1.8V供电电压,功耗为3.6mW,1MHz频点附近的噪声系数约为17dB,1dB输入压缩点为14.6dBm。     

A mismatch-error minimized four-channel time-interleaved 11 b 150 MS/s pipelined SAR ADC

This work proposes a four-channel time-interleaved 11 b 150 MS/s pipelined SAR ADC based on various analog techniques to minimize mismatches between channels without any calibration scheme. The proposed ADC eliminates an input SHA to reduce offset mismatches, while the pipelined SAR architecture solves the problem of limited input bandwidth as observed in conventional SHA-free ADCs. In addition, a shared residue amplifier between four channels minimizes various mismatches caused by amplifiers in the first-stage MDACs. Two types of references for the residue amplifier and the SAR ADCs isolate the reference instability problem due to different functional requirements, while the shared residue amplifier uses only a single reference during the amplifying mode of each channel to reduce a gain mismatch. For high performance of the SAR ADC, high-frequency clocks with a controllable duty cycle are generated on chip without external, complicated, high-speed multi-phase clocks. The prototype 11 b ADC in a 0.13 μm CMOS shows a measured DNL and INL of 0.31 LSB and 1.18 LSB, respectively, with an SNDR of 59.3 dB and an SFDR of 67.7 dB at 100 MS/s, and an SNDR of 54.5 dB and an SFDR of 65.5 dB at 150 MS/s. The ADC with an active die area of 2.42 mm² consumes 46.8 mW at 1.2 V and 150 MS/s.     

Sub 1-V highly-linear low-power class-AB bulk-driven tunable CMOS transconductor

This paper describes the design and realization of a sub 1-V low power class-AB bulk-driven tunable linear transconductor using a 0.18-μm CMOS technology. The proposed transconductor employs a class-AB bulk-driven differential input voltage follower and a passive resistor to achieve highly linear voltage-to-current conversion. Transconductance tuning is achieved by tuning the differential output current of the core transconductor with gain-adjustable current mirrors. With 10.4-μA current consumption from a 0.8-V single power supply voltage, simulation results show that the proposed transconductor achieves the total harmonic distortion (THD) of <?40 dB for a peak differential input voltage range of 800 mV at frequencies up to 10 kHz. The simulated input-referred noise voltage integrated over 10-kHz bandwidth is 100 μV, resulting to an input signal dynamic range of 75 dB for THD <?40 dB. A biquadratic G_m-C filter is designed to demonstrated the performance of the proposed transconductor. At the nominal 10-kHz cut-off frequency, the filter dissipates 34.4 μW from a 0.8-V supply voltage and it achieves an input signal dynamic range of 67.4 dB for the third-order intermodulation distortion of <?40 dB.     

System simulations of a 1.5 V SiGe 81–86 GHz E-band transmitter

This paper presents simulation results for a sliding-IF SiGe E-band transmitter circuit for the 81–86 GHz E-band. The circuit was designed in a SiGe process with f _T  = 200 GHz and uses a supply of 1.5 V. The low supply voltage eliminates the need for a dedicated transmitter voltage regulator. The carrier generation is based on a 28 GHz quadrature voltage oscillator (QVCO). Upconversion to 84 GHz is performed by first mixing with the QVCO signals, converting the signal from baseband to 28 GHz, and then mixing it with the 56 GHz QVCO second harmonic, present at the emitter nodes of the QVCO core devices. The second mixer is connected to a three-stage power amplifier utilizing capacitive cross-coupling to increase the gain, providing a saturated output power of +14 dBm with a 1 dB output compression point of +11 dBm. E-band radio links using higher order modulation, e.g. 64 QAM, are sensitive to I/Q phase errors. The presented design is based on a 28 GHz QVCO, the lower frequency reducing the phase error due to mismatch in active and passive devices. The I/Q mismatch can be further reduced by adjusting varactors connected to each QVCO output. The analog performance of the transmitter is based on ADS Momentum models of all inductors and transformers, and layout parasitic extracted views of the active parts. For the simulations with a 16 QAM modulated baseband input signal, however, the Momentum models were replaced with lumped equivalent models to ease simulator convergence. Constellation diagrams and error vector magnitude (EVM) were calculated in MATLAB using data from transient simulations. The EVM dependency on QVCO phase noise, I/Q imbalance and PA compression has been analyzed. For an average output power of 7.5 dBm, the design achieves 7.2% EVM for a 16 QAM signal with 1 GHz bandwidth. The current consumption of the transmitter, including the PA, equals 131 mA from a 1.5 V supply.     

Wideband MOSFET differential V/I converter

This article presents a proposed modified flipped voltage follower for applications in a wideband MOSFET differential V/I converter using IBM 0.13 μm technology. Operating with ±2.5 V supply rails, the transconductance is nominally 3.3 mS for an input differential signal voltage range of 0.5 V, and the ?3 dB bandwidth exceeds 4 GHz. The THD measured at 1 MHz for a differential input signal of 500 mVp-p is less than –82 dB, and is still below –50 dB at 1 GHz.     

A 75-dB image rejection IF-input quadrature-sampling SC /spl Sigma//spl Delta/ Modulator

Quadrature sampling of intermediate frequency (IF) signals is subject to the well-known problem of gain and phase mismatches between the in-phase (I) and quadrature (Q) channels. This paper presents an IF-input quadrature-sampling switched-capacitor (SC) /spl Sigma//spl Delta/ modulator that circumvents the I/Q mismatch problem by time-sharing between the I and Q channels the critical circuit components, namely, the sampling capacitor and the capacitor of the first-stage feedback digital-to-analog converter (DAC). In addition, a clocking scheme that is insensitive to I/Q phase imbalance is used. A third-order single-loop 1-bit low-pass modulator has been designed and fabricated in a 0.35-/spl mu/m CMOS process with an active area of 0.57mm/sup 2/. The experimental results show that the modulator achieves an image-rejection ratio (IRR) of greater than 75dB throughout a 200-kHz signal bandwidth.     

A SAR-based ΣΔ modulator using shared DAC

In this paper a new successive approximation (SA) quantizer based on the elimination of the digital to analog converter (DAC) from the quantizer structure is presented. Instead; the feedback DAC block of the ΣΔ modulator is shared by SA quantizer. Using an efficient decoding algorithm in the proposed structure in conjunction with the above SA quantizer DAC elimination method, results in a reduction of the level number of the feedback DAC, and hence, a significant drop in power and area consumption is achieved. In order to study the performance of the proposed structure, a third order discrete-time ΣΔ modulator is designed and simulated in 0.18 μm CMOS technology with the following performance characteristics; a signal to noise ratio of 79.2 dB, dynamic range of 84.8 dB, power consumption of 3.75 mW and a figure of merit of 0.66 pJ/conv from a 1.8 V supply with an input signal of 200 kHz bandwidth.     

Preliminary Design of a Harmonic-doubling Gyrotron Traveling Wave Amplifier

This study proposes a Ka-band harmonic-doubling gyrotron traveling-wave amplifier (gyro- TWT), using distributed wall losses in the input stage and mode-selective interaction circuit in the output stage, to improve the stability of the amplification. Based on a large signal simulation code, a saturated peak power of 163 kW with an efficiency of 15.5%, a gain of 31.1 dB, and a 3 dB bandwidth of 0.9 GHz is predicted for the gyro-TWT driven by 70 kV, 15 A electron beam with a velocity ratio of 1.2 and velocity spread 5% at 33.2 GHz.     

Zero-IF topology

A new receiver topology called the double-quadrature zero-IF topology is presented. Studies have shown that with the new topology the sensitivity to phase errors of quadrature channels can be significantly decreased, and that the image-recognition ratio (IRR) against the gain mismatch characteristic is inherently superior to that for the conventional zero-IF scheme by ~6 dB     

A Filter Bank Mismatch Calibration Technique for Frequency-Interleaved ADCs

The filter bank mismatch of analog analysis filters in frequency-interleaved ADCs (FI-ADCs) degrades the system’s spurious-free dynamic range (SFDR) significantly. In this paper, a calibration approach for compensating such mismatch is presented. By modeling the parameter mismatches in the analysis filters, the filter bank mismatch compensation is divided into a coarse trimming mode and a fine-tuning mode. After the coarse trimming mode by trimming the resistors and capacitors in analog domain, the fine-tuning mode by updating coefficients of synthesis filters is further carried out in digital domain to achieve high-precision calibration. A design example of 10 GS/s 8-bit four-channel FI-ADC is built in MATLAB. The simulation results show that 25-tap synthesis filters could satisfy the reconstruction requirement of 8-bit ADC. The proposed calibration technique improves the SFDR to 51 dB, compensating the filter mismatch effectively.     

A 60 GHz receiver front-end with PLL based phase controlled LO generation for phased-arrays

This paper presents a front-end architecture for fully integrated 60 GHz phased array receivers. It employs LO-path beamforming using a phase controlled phase-locked loop (PC-PLL). To demonstrate the architecture a circuit is implemented featuring a two stage low noise amplifier, two cascaded active mixers, and a PC-PLL. The receiver downconverts the 60 GHz signal in two steps, using LO signals from the 20 GHz QVCO of the PLL. A differential 2nd-order harmonic is coupled from the sources of the current commutating pairs of the QVCO, feeding the LO-port of the first mixer and downconverting the 60 GHz RF signal to a 20 GHz intermediate frequency. Quadrature 20 GHz LO signals are then used in the second mixer to down-convert the IF signal to baseband. The PLL is locked to a relatively high reference frequency, 1.25 GHz, which reduces the size of the PLL loop filter and enables a compact layout. The measurements show an input return loss better than ?10 dB between 57.5 and 60.8 GHz, a 15 dB voltage gain, and a 9 dB noise figure. Two-tone measurements show ?12.5 dBm IIP3, 29 dBm IIP2, and ?24 dBm ICP1. The PC-PLL phase noise is ?105 dBc/Hz at 1 MHz offset from a 20 GHz carrier, and the phase of the received 60 GHz signal is digitally controllable with a resolution of 3.2°, covering the full 360° range with a phase error smaller than 1°. The chip consumes 80 mA from a 1.2 V supply, and measures 1,400 μm × 660 μm (900 μm × 500 μm excluding pads) including LNAs, mixers, and PC-PLL in a 90 nm RF CMOS process.